Power inverter for feeding electric energy from a dc power generator into an ac grid with two power lines

ABSTRACT

A power inverter includes two input terminals, two output terminals and a resonant converter that includes a high frequency transformer having a primary winding and a secondary winding, at least one high frequency switched semiconductor power switch that connects one end of the primary winding of the high frequency transformer to one of the input terminals to provide a current path through the primary winding to the other one of the input terminals. The power inverter further includes a resonant series circuit having an inductance and a capacitance, and a high frequency rectifier that rectifies a current through the secondary winding, two output lines, and an output converter connected between the output lines of the high frequency rectifier and the two output terminals.

REFERENCE TO RELATED APPLICATIONS

This application is a continuation of International application numberPCT/EP2011/070011 filed on Nov. 14, 2011, which claims priority toInternational application number PCT/EP2010/067355, filed on Nov. 12,2010.

FIELD

The present disclosure generally relates to a power inverter for feedingelectric energy from a DC power generator into an AC grid with two-powerlines. Further, the present disclosure relates to a method of operatingsuch a power inverter.

BACKGROUND

In some known power inverters for feeding electric energy from a DCpower generator into an AC grid with two-power lines, the power invertercomprises a DC/DC converter for matching the output voltage of the powergenerator to the grid, and a DC/AC output converter for actually feedingthe electric power from the DC power generator into the AC grid.

In such known power inverters the DC/DC converter can be a converterincluding a high frequency transformer comprising a primary winding anda secondary winding. Such a transformer generally provides for agalvanic isolation of the secondary or output side from the primary orinput side of the power inverter. A DC/DC converter including a highfrequency transformer further comprises at least one high frequencyswitched semiconductor power switch that, in its closed state, connectsone end of the primary winding of the high frequency transformer to oneof the input terminals of the power inverter for providing a currentpath through the primary winding to the other one of the inputterminals. The alternating current through the primary winding may alsobe provided by any type of inverter bridge connected between the inputterminals of the power inverter.

The current through the secondary winding of the high frequencytransformer is rectified by a high frequency rectifier typicallycomprising diodes arranged as a rectifier bridge, and a filter capacitorconnected between output lines of the high frequency rectifier.

An interesting sub-class of DC/DC converters used in known powerinverters are resonant or quasi-resonant converters, which comprise aresonant circuit. Such a resonant circuit allows for zero voltage and/orzero current switching of the semiconductor power switches providing thealternating current through the primary winding of the high frequencytransformer. The resonant circuit may be provided on the primary side oron the secondary side of the high frequency transformer, and it can be aresonant parallel or series circuit.

Bob Mammano and Jeff Putsch: Fix-Frequency, Resonant-Switched PulseWidth Modulation with Phase-Shifted Control(http://server.oersted.dtu.dk/ftp/database/Data_CDs/component_data/Unitrode_seminars/se_m800/slup096.pdf)disclose a resonant-switched DC/DC converter comprising two half bridgesconnected between DC input terminals, each half bridge comprising twosemiconductor switches and a center connected to one respective end of aprimary winding of a high frequency transformer. The ends of thesecondary winding of the high frequency transformer are each connectedto a rectifier diode. One end of a filter circuit comprising a seriesinductor and an output capacitor is connected to a joint output of bothrectifier diodes, and its other end is connected to a center tap of thesecondary winding. The semiconductor power switches of each half bridgeat the primary side of the high frequency transformer are controlled bycomplementary high frequency signals of 50% duty cycle. Thus, at anytime at least one of the semiconductor switches of each half bridge isclosed except for a dead time during which the parallel resonanttransition occurs. If the semiconductor switches of the two half bridgesconnected to the same input terminal are closed at the same time, theprimary winding of the high frequency transformer is short-circuited viathese two semiconductor power switches. Only if just one of theseswitches is closed whereas the other is open, a current between theinput terminals flows through the primary winding of the high frequencytransformer. These on-times of the primary side of the high frequencytransformer are defined by a phase-offset or phase-shift between thehigh frequency signals applied to both half bridges. The length of theon-times is defined with conventional PWM and controls the powerdelivered to the load. Switching of the semiconductor power switches isdone at zero voltage.

Both zero voltage switching and its dual equivalent, zero currentswitching, provide for very low switching losses. However, in zerocurrent switching, it is not possible to use pulse width modulation as acurrent shaping means with high efficiency. Only a modulation of therepetition rate of the pulses is available, as the pulse widths aredetermined by the zero current switching criterion.

The DC/AC converters at the output side of some known power invertersare inverter bridges with high frequency switched semiconductor powerswitches for forming a desired sine shape of the currents fed into theAC grid. Some other known power inverters, however, comprise aline-commutated converter at their output end, the switching elements ofwhich are only controlled by the voltages of a connected AC grid, andare, thus, only switching at the grid frequency. As a result, theseDC/AC converters at the output side are unable to provide a sine shapeof the current fed into the AC grid, and any current shaping has to bedone upstream of the line-commutated converter.

Some photovoltaic modules show a premature degradation in use if notpermanently operated at a negative or positive electric potential withregard to electric ground. Further, operating photovoltaic modules at adefined negative or positive electric potential could be used for groundfault detection. Thus, some efforts are made to provide a voltage offsetfor the input terminals of a power inverter for feeding electric energyfrom such photovoltaic generators into an AC grid.

Some known power inverters require particular attention during theirstartup, as dangerously high currents may flow as long as buffercapacitors are not yet loaded to provide a sufficient counter voltage.On the other hand, electric loads present on buffer capacitors inoperation of some known power inverters pose a danger when terminatingthe operation of known power inverters, even if all active parts of thepower inverters have been inactive for some time and even if the powerinverter has already been disconnected from the grid for some time.

Some regulations, like those in the US, require a galvanic isolationfrom the public power grid for any power generator from which electricpower is fed into the public power grid.

A power inverter for feeding electric energy from a DC power generatorinto an AC power grid with two-power lines is known from DE 10 2005 023290 A1. This power inverter is a bi-directional battery inverter andcomprises a high frequency transformer. The high frequency transformerand a resonant capacity connected to the secondary winding of the highfrequency transformer form a resonant series circuit. The primarywinding of the transformer has a center tap and is connected to thebattery via a center point circuitry with semiconductor switches. Theresonant series circuit is connected to a rectifier. The rectifier isconnected to a boost converter that feeds a DC input voltage link of aDC/AC converter.

DE 10 2005 023 291 A1 discloses a power inverter including a resonantconverter. The resonant converter comprises a high frequency transformerthat forms a resonant series circuit in combination with a resonantcapacitance connected to its primary winding. The secondary winding ofthe high frequency transformer is connected to a rectifier that isconnected to a DC voltage input link of a DC/AC converter.

US 2008/0192510 A1 discloses a power inverter similar to the one knownfrom DE 10 2005 023 291 A1. Here, the primary winding of the highfrequency transformer is fed by a photovoltaic generator by means of aninverter full bridge, the center point of each of the half bridges ofthe inverter full bridge being connected to one end of the primarywinding.

U.S. Pat. No. 5,587,892 A discloses a multi-phase power converter withharmonic neutralization, in which capacitors are connected to each endof a primary winding of a high frequency transformer. Each of thesecapacitors is combined with an inductor in addition to the highfrequency transformer to provide a resonant series circuit.

There still is a need for a power inverter particularly suitable forfeeding electric energy from small to medium sized photovoltaic modulesinto an AC power grid, the inverter being available at low cost butnevertheless displaying a high performance, i.e. low power losses at ahigh level of security.

SUMMARY

The present disclosure relates to a power inverter for feeding electricenergy from a DC power generator into an AC grid with two-power lines.The inverter comprises two input terminals for connecting the powergenerator, two output terminals for connecting the two power lines ofthe AC grid, and a resonant converter including a high frequencytransformer comprising a primary winding and a secondary winding, and atleast one high frequency switched semiconductor power switch. In itsclosed state the semiconductor power switch connects one end of theprimary winding of the high frequency transformer to one of the inputterminals for providing a current path through the primary winding tothe other one of the input terminals. The resonant converter furtherincludes a resonant series circuit comprising an inductance and acapacitance, a high frequency rectifier rectifying a current through thesecondary winding of the high frequency transformer and having twooutput lines, and an output converter connected between the output linesof the high frequency rectifier and the two output terminals.

In one embodiment of the present disclosure, the resonant converterfurther comprises a controller that is connected to the output terminalsfor receiving a voltage signal and provides high frequency controlsignals for controlling all semiconductor power switches of the resonantconverter to sine-modulate an AC current fed into the AC grid in phasewith the voltage signal. This means that the sine-modulation isperformed by appropriately controlling the resonant converter instead ofthe output converter, which thus just feeds the half-waves of thecurrent that have already been sine shaped to the correct outputterminals. Particularly, the controller may vary a repetition rate ofpulses in the high frequency control signals for sine-modulating the ACcurrent fed into the AC grid. Further, the controller may vary anaverage repetition rate of the pulses in the high frequency controlsignals for controlling the electric power fed from the power generatorinto the AC grid.

In a more detailed embodiment of the present disclosure the resonantconverter further comprises two half bridges connected between the inputterminals, wherein each half bridge has two high frequency switchedsemiconductor power switches and a center. A primary winding of the highfrequency transformer is connected between the centers of the two halfbridges, and the controller provides one high frequency control signalper half bridge that is controlling (e.g., directly controlling) one ofthe two semiconductor power switches of the respective half bridge andthat is inversed for controlling the other of the two semiconductorpower switches of the respective half bridge for zero current switching(ZCS) of the semiconductor power switches of the half bridges. In thisaspect, the controller particularly provides the high frequency controlsignals for full wave mode (FWM) zero current switching of thesemiconductor power switches of the half bridges, wherein the controllervaries a length of on-times of the primary winding, during which one ofthe semiconductor power switches of one of the half bridges that isconnected to one of the input terminals and one of the semiconductorpower switches of the other of the half bridges that is connected to theother of the input terminals are closed while the two othersemiconductor power switches of the half bridges are open, in a rangefrom about 50% to about 100% of the resonance period. Even moreparticularly, the controller may provide the two control signals at adelay or time shift with partially overlapping pulses, the partiallyoverlapping pulses in both control signals being of equal length that isat least as long as the resonance period of the resonant series circuitof the resonant converter, and may vary the time shift between the twohigh frequency signals for varying the length of the on-times.

In a further more detailed embodiment of the present disclosure, thepower inverter further comprises an electrical isolation barrier betweenthe input terminals and the output terminals, the high frequencyswitched semiconductor power switches and the primary winding of thehigh frequency transformer being on the same side of the barrier as theinput terminals, and the resonant series circuit including the secondarywinding of the high frequency transformer, the high frequency rectifierand the output converter being on the same side of the barrier as theoutput terminals, and all lines crossing the isolation barriercomprising at least one of a high ohmic resistor or a capacitor at theisolation barrier.

In a further embodiment of the present disclosure, the resonantconverter of the power inverter further comprises two half bridgesconnected between the input terminals, wherein each half bridge has twohigh frequency switched semiconductor power switches and a center. Aprimary winding of the high frequency transformer is connected betweenthe centers of the two half bridges. A method of operating this powerinverter comprises generating two high frequency control signals at atime shift between the two high frequency control signals, and applyingthe two high frequency control signals to those two semiconductor powerswitches of the two half bridges that are connected to one of the twoinput terminals to provide for on-times in which only one of these twosemiconductor power switches is closed. In one embodiment, a length ofthe on-times is in a range from 50% to 100% of a resonance period of theresonant series circuit of the resonant converter. The method furthercomprises generating two inverse high frequency control signals, eachbeing the inverse of one of the two high frequency control signals, andapplying the two inverse high frequency control signals to those twosemiconductor power switches of the two half bridges that are connectedto the other of the two input terminals, such that each high frequencycontrol signal and the inverse of that control signal are applied to thetwo semiconductor power switches of the same half bridge. Finally, themethod comprises synchronously modulating all high frequency controlsignals to sine-modulate the AC current fed into the AC grid in phasewith a voltage of the AC grid.

In a further embodiment of the present disclosure, which is also ofimportance for other types of power inverters, at least one varistorprovided in an EMC-filter connected between the output converter and thetwo output terminals. The embodiment further comprises chokes selectedfrom common mode chokes and combi mode chokes that are connected betweencenter taps or other suitable taps of the common mode chokes or combimode chokes. Combi mode chokes are sometimes also referred to as “hybridchoke coils”. Their actual design will be described later.

In a further embodiment of the present disclosure, the power invertercomprises a ground terminal for connecting a power grid ground line,wherein one of the input terminals is connected to one of the inputlines of the line-commutated full bridge output converter via a resistoron the one hand, and to the ground terminal via a capacitor on the otherhand. As a result, the one input terminal exhibits a fixed potentialoffset that is adjustable between electric ground and the averagevoltage present at the connected one of the output lines of the highfrequency rectifier by the ratio of a resistance of the resistorconnecting the input terminal to the input line of the output converterand of a resistance of a further resistor connected in parallel to thecapacitor connecting the input terminal to the ground terminal.

Other features and advantages of the present disclosure will becomeapparent to one with skill in the art upon examination of the followingdrawings and the detailed description. It is intended that all suchadditional features and advantages be included herein within the scopeof the present disclosure, as defined by the claims.

BRIEF DESCRIPTION OF THE DRAWINGS

The disclosure can be better understood with reference to the followingdrawings. In the drawings, like reference numerals designatecorresponding parts throughout the several views.

FIG. 1 is a circuit diagram of a first example of the power inverterapplicable for feeding electric power from a photovoltaic powergenerator into a single phase AC grid according to the Europeanstandard.

FIG. 2 is a circuit diagram of the same topology as depicted in FIG. 1indicating that it is also applicable for feeding electric power fromthe photovoltaic power generator or any other DC power generator into asplit-phase AC grid according to the US standard.

FIG. 3 shows combi mode chokes, which may be used in an EMC-filter ofthe power converter according to FIG. 1 or 2.

FIG. 4 illustrates the connection of a varistor to taps of common modechokes or combi mode chokes for use in an EMC-filter of the powerinverter of FIG. 1 or 2.

FIG. 5 is a timing diagram of one possible embodiment of high frequencycontrol signals applied to semiconductor power switches of the powerinverter of FIG. 1 or 2.

FIGS. 6 to 10 illustrate a current through a primary winding and avoltage across a secondary winding of a high frequency transformer ofthe power inverter of FIG. 1 or 2 at a constant input voltage providedby a connected power generator and at different instantaneous outputvoltages defined by a connected AC grid.

FIG. 11 is a timing diagram of another possible embodiment of highfrequency control signals for the semiconductor power switches of thepower inverter according to FIG. 1 or 2.

FIG. 12 illustrates a spatial arrangement according to one embodiment ofthe semiconductor power switches controlled by the high frequencysignals according to FIG. 11.

FIG. 13 shows a plurality of photovoltaic modules connected to the ACgrid via a plurality of power inverters, a ring of power lines and aconnector box.

DETAILED DESCRIPTION

In one embodiment, the DC power generator may be a photovoltaic powergenerator. Even more particular, the photovoltaic power generator may bea small to medium sized photovoltaic module, and the power inverter mayonly be provided for feeding the electric energy from this photovoltaicmodule into the AC grid.

The AC grid with two-power lines may be a single phase AC grid in whichone of the two power lines is a phase line whereas the other power lineis a neutral line. Such an AC grid may in particular be a single phaseAC grid according to the European standard. The AC grid with two-powerlines may also be a split-phase grid with a neutral midpoint, thetwo-power lines being the two wires between which the split-phasealternating voltage is present. Such an AC grid may in particular be asplit-phase grid according to the US standard.

Referring now in greater detail to FIGS. 1 and 2, a power inverter 1comprises two input terminals 2 and 3, two output terminals 4 and 5, anda ground terminal 6. The two input terminals 2 and 3 are provided forconnecting a DC power generator, like, for example, a photovoltaic powergenerator. In one embodiment, the power generator is a photovoltaicmodule providing a DC output voltage of 12 to 60 Volts, like, forexample, about 25 Volts. The output terminals 4 and 5 and the groundterminal 6 are provided for connecting the power inverter 1 to an ACpower grid having two power lines to be connected to the outputterminals 4 and 5 and a ground line to be connected to the groundterminal 6. The peak voltage between the power lines of the AC grid,which is applied between the output terminals 4, 5, may be up to about400 V. The power inverter 1 feeds electric power from the powergenerator connected to the input terminals 2 and 3 into the AC gridconnected to the output terminals 4 and 5.

The electric potentials PV+ and PV− applied to the input terminals 2 and3 are applied to two inverter half bridges 7 and 8. Each inverter halfbridge 7 and 8 comprises two semiconductor power switches 9 and 10, and11 and 12, respectively. If the semiconductor power switches 9 to 12 areMOSFETs as diagrammatically indicated here, they each include ananti-parallel diode 13. The anti-parallel diodes 13 are, however, notrequired for the function of the power inverter 1 that is describedhere. Further, each half bridge 7 and 8 has a center 14 and 15,respectively. A primary winding 16 of a high frequency transformer 17 isconnected between the center points 14 and 15 of the half bridges 7 and8, and the semiconductor power switches 9 to 12 of the half bridges 7and 8 are controlled to conduct current from the power generatorconnected to the input terminals 2 and 3, which is not depicted here, inalternating directions through the primary winding 16. When no currentshall be conducted from the power generator connected to the inputterminals 2 and 3 through the primary winding, the semiconductor powerswitches 9 to 12 of the half bridges 7 and 8 are controlled toshort-circuit the ends of the primary winding 16.

When all semiconductor power switches 9 to 12 of the half bridges 7 and8 are open, the power generator that is connected to the input terminals2, 3 is effectively disconnected from the AC grid that is connected tothe output terminals 4 and 5. Thus, there is no need for an additionalrelay to provide this separation. Even if one of the semiconductor powerswitches does not open due to a failure, the power generator would stillbe separated from the AC grid. This means a higher level of securitythan with a relay, the contacts of which may weld together. If for somereason an additional relay is to be provided between the power inverterand the AC grid, this relay may be provided for a plurality of powerinverters each feeding electric power from one power generator into theAC grid. This aspect of the present disclosure will be explained in moredetail later, with reference to FIG. 13.

The high frequency transformer 17 is part of a resonant series circuit19 that further comprises two capacitors 20 and 21 of equal capacitance.The capacitors 20 and 21 are symmetrically connected to a secondarywinding 18 of the high frequency transformer 17 with one of their ends,i.e., one end of each capacitor is connected to each end of thesecondary winding 18. There is no separate inductor besides the highfrequency transformer 17 providing the inductance of the resonant seriescircuit 19. Thus, the losses due to magnetic leakage are kept to aminimum here. The provision of the capacitance of the resonant seriescircuit 19 by two capacitors 20 and 21 that are connected to both endsof the primary winding 18 reduces the required electric strength forboth the capacitors and the high frequency transformer, and reducescommon mode injection of noise and therefore requires less filtering.Further, it is possible to use Y-capacitors here in one embodiment.

A high frequency rectifier 22 designed as a rectifier full bridge 23 ofdiodes 24 rectifies the current in the resonant series circuit 19 on thesecondary winding 18 side of the high frequency transformer 17 thatresults from the voltage induced by the current through the primarywinding 16. A capacitor 25 connected between output lines 26 and 27 ofthe high frequency rectifier rejects the high frequency components ofthe rectified current but lets the low frequency components of thisrectified current pass. Particularly, it has a cut-off frequency above agrid frequency of the AC grid connected to the output terminals 4 and 5.For practical reasons the cut-off frequency may be in the order of a fewkHz. The resulting small capacitance of the capacitor 25 allows for acapacitor 25 of low dimensions and little cost; it also ensures thatonly very little reactive power is drawn by the capacitor 25 out of theAC grid connected to the output terminals 4 and 5 via an outputconverter 31, and it neither excites dangerously high currents into thecapacitor 25 during a start-up, nor from the capacitor 25 after stoppingand disconnecting the power inverter from the AC grid. The parts of thepower inverter 1 between the input terminals 2 and 3 and the outputlines 26 and 27 described up to know constitute a resonant converter 51.The capacitor 25, however, may also be placed further downstream, like,for example, at the output end of the output converter 31.

The differential voltages present at the output lines 26 and 27 as wellas between the output lines 26 and 27 during one period of the gridfrequency are indicated in small V(t) plots 28, 29 and 30, respectively.While the differential voltages present at the output lines 26 and 27 donot differ between FIGS. 1 and 2, the variation of the voltages in timeon the output lines 26 and 27 depend on the kind of grid connected tothe output terminals 4 and 5. This is particularly the case as theoutput lines 26 and 27 of the resonant converter 51 are connected to theoutput terminals 4 and 5 via the line-commutated full bridge converter31 that serves as an unfolding bridge correctly unfolding the currenthalf waves flowing through the output lines 26 and 27 to the outputterminals 4 and 5. Such line-commutated converters, in which switchingelements of the full bridge are only controlled by the voltages of theexternal AC grid connected to the output terminals 4 and 5, aregenerally known.

It is also generally known to have an EMC-filter 32 connected betweenthe output converter 31 and the output terminals 4 and 5 to care forelectromagnetic compliance. However, the EMC-filter 32 may deviate froma standard EMC-filter in that it comprises common mode chokes or in oneembodiment combi mode chokes instead of standard chokes. Common modechokes are arranged on a common magnetic core that typically is in aring shape. Common mode chokes have to be combined with standard chokesarranged between the common mode chokes and the output terminals. Combimode chokes 56, an example of which is depicted in FIG. 3, are arrangedon a common ring magnetic core 57 that has an additional web 58, forexample, comprising an air gap 59, and do not require additional chokeswithin the EMC-filter 32. An electrically conductive material may beprovided in the air gap 59 to achieve an increased attenuation of highfrequency oscillations of the currents flowing through the chokes. Thisattenuation is due to eddy currents generated in the electricallyconductive material by such oscillations. Suitable electricallyconductive materials are copper and aluminum, which are, for example,provided as a foil extending through the air gap 59.

Both common mode chokes and combi mode chokes 56 of the EMC-filter 32are particularly well suited for the connection of a varistor 60protecting the power inverter 1 against overvoltages in the AC grid thatis connected to the output terminals 4, 5, and vice versa protecting theAC grid against overvoltages occurring in the power inverter 1. FIG. 4shows how the varistor 60 is connected between taps 61 of the two combimode chokes 56. A further varistor, not depicted here, may be addedbetween the combi mode chokes 56 at their input ends at the left handside of FIG. 4. In case of an overvoltage 62 that may come from theright hand side of the combi mode chokes 56 depicted in FIG. 4 and thatcauses a current through the varistor 60, the parts of the common modechokes on the right hand side of the taps 61 in FIG. 4 choke the currentthrough the varistor 60. At the same time an additional voltage drop inthe parts of the combi mode choke 56 on the left hand side of the taps61 in FIG. 4 is induced that coincides with the voltage drop due to theactual current through the varistor 60, because of the magnetic couplingof the combi mode choke 56 via the ring magnetic core 57. As a result,the varistor needs to get conductive only at comparatively highovervoltages 62 and has to stand only comparatively low currents butnevertheless effectively avoids the transmittance of overvoltages acrossthe combi mode chokes 56. A varistor connected to taps of common mode orcombi mode chokes is not only of advantage in a power inverter 1 asdescribed here. This combination of a varistor and common mode or combimode chokes may also be used in EMC-filters of other power inverters orin other technical fields.

Finally, fuses 33 are connected between the EMC-filter 32 and theindividual output terminals 4 and 5 of the power inverter according toFIG. 1 or 2. Small V(t) plots 34 and 35 indicate the variations of thevoltages in time applied by an external AC grid to the output terminals4 and 5, which by means of the line-commutating converter 31 define thevariations of the voltages in time indicated in the V(t) plots 28 and 29on the output lines 26 and 27 of the resonant converter 51 depending onthe variation of the voltage in time between the output lines 26 and 27indicated in the V(t) plot 30, which is determined by the operation ofthe half bridges 7 and 8. However, independent of the standard of the ACgrid connected to the output terminals 4 and 5, the average voltage onoutput line 26 is always positive and the average voltage on output line27 is always negative with regard to the neutral line N connected tooutput terminal 5 in FIG. 1 or to the midpoint neutral between the linesL1 and L2 connected to the output terminals 4 and 5 in FIG. 2. Thevalues of theses average voltages are both at about 50% of the effectivevalue of the voltage applied between the output terminals 4 and 5. Thenegative average voltage on line 27 is used to provide a negative offsetvoltage to the input terminal 3, here. To this end, line 27 is connectedto input terminal 3 via two resistors 36 and 37, which could be replacedby a singe resistor but which is not done here on purpose, as explainedlater. Further, input terminal 3 is connected to ground connector 6 viaa parallel connection of another resistor 38 and a capacitor 39 and,thus, connected to the ground of the AC grid, which is connected to theground connector 6. In principle, the resistor 38 in this connection ofinput terminal 3 to ground is optional. Here, the resistors 36 to 38provide a voltage divider defining a voltage offset for input terminal 3that is at a defined voltage level between electrical ground and theaverage voltage on line 27. This offset voltage of the input terminal 3does not exhibit the time course depicted in plot 29 but is temporallyaveraged by capacitor 39. Still the capacitor 39 is of small capacitanceonly to ensure a connection of the input terminal 3 to ground for highfrequency currents and voltages for security reasons. If the absolutevalue of the negative offset voltage at the input terminal 3 is higherthan the output voltage of the photovoltaic power generator connected toinput terminals 2 and 3, this allows for operating the photovoltaicpower generator completely at a negative potential with regard toground, which has huge advantages with some photovoltaic powergenerators for avoiding premature degradation.

A capacitor 40 that has a high capacitance is connected between theinput terminals 2 and 3 and serves as a buffer, stabilizing the voltagedifference between the input terminals 2 and 3, which is provided by theconnected power generator, during pulsed operation of the half bridges 7and 8. The capacitor 40 is particularly used to suppress the 100 or 120Hz ripple voltage that is characteristic for single phase inverters.

The two half bridges 7 and 8 of the resonant converter 51 are operatedby a controller 41 via drivers 52 and 53. Generally, the controller 41operates the half bridges 7 and 8 in such a way that the current flowingthrough lines 26 and 27 consists of sine-shaped half waves that are inphase with the AC grid connected to the output terminals 4 and 5. Noshaping of the current fed into the AC grid by the power inverter 1 ispossible by the output converter 31 as long as it is line-commutated.The controller 41 receives the voltages applied by the AC grid to theoutput terminals 4 and 5 via signal lines 42 and 43, in which isolationresistors 44 and 45 are arranged. The resistors 44 and 45 in the signalline 42 are bypassed by a capacitor 46 to enable the controller 41 toreceive high frequency power line communication signals via the signalline 42 without attenuation by the resistors 44 and 45. The linefrequency signals Va and Vb received by the controller 41 via the signallines 42 and 43 are used for synchronizing the operation of the halfbridges 7 and 8 with the alternating voltage of the AC grid connected tothe output terminals 4 and 5. The controller 41 provides two pulsedcontrol signals R and L and an enable signal E to the drivers 52 and 53via control lines 47 to 49. The enable signal turns on and off thedrivers 52 and 53 and, thus, the entire power inverter 1.

According to one possible embodiment illustrated in FIG. 5, the controlsignal R consists of pulses 63 that typically have a fixed duty cycle ofabout 50%. The repetition rate of the pulses 63 is variable up to abouthalf the resonance frequency of the resonant series circuit 19. Thus,the duration of the pulses is at least as long as the resonance periodof the resonant series circuit 19. The control signal L is equal to thecontrol signal R, but time-shifted by 50 to 100% of the resonance periodof the resonant series circuit 19. The driver 52 inverts the controlsignal L for operating the semiconductor switch 9 and uses it directlyfor operating the semiconductor switch 10. Likewise, the driver 53inverts the control signal R for operating the semiconductor switch 11and uses it directly for operating the semiconductor switch 12. Theinverted signals L, inv. and R, inv. are depicted at the bottom of FIG.5. Thus, always one of the switches 9 and 10 and one of the switches 11and 12 is closed, whereas the other is open. During the transientsbetween the switches 9 and 10 and the switches 11 and 12, respectively,a small dead time may be inserted during which none of the two switchesis closed. This means that the signals L, inv. and R, inv. may somewhatdiffer from a true inversion of the signals L and R. However, such adead time will always be much smaller than the on-time of the switches.For example, the dead time may be 20 ns, and the on-time may be 0.5 us.The half bridges 7 and 8 operated in this way only provide a currentpath between the input terminals 2 and 3 through the primary winding 16of the high frequency transformer 17 if the values of the controlsignals R and L differ from each other. At other times than theseon-times, i.e. when the signals R and L have the same values, theprimary winding 16 is either short-circuited by the switches 9 and 11 orthe switches 10 and 12, which are both closed then and, thus, connectthe center points 14 and 15 during these times. Varying the temporaloverlap and, thus, inversely proportionally varying the length of theon-time, during which the primary winding 16 is connected to the inputterminals 2 and 3, in a range from 50% to 100% of the resonance periodof the resonant series circuit 19 allows for full wave mode zero currentswitching of the switches 9 to 12 despite of a variation of the point intime of the second zero crossing of the current after the start of theon-time. The optimum length of the on-times for the primary winding 16and the corresponding optimum overlap of the pulses 63 of the controlsignals R and L primarily depends on the voltage applied by the AC gridvia the output converter 31, which strongly influences the time-courseof the current through the primary winding 16 during the second half ofthe resonance period of the series resonant circuit 19. This will beexplained in detail later with reference to FIGS. 6 to 10.

With full wave mode zero current switching it is not possible to formthe time-dependent shape of the currents flowing through lines 26 and 27by pulse width modulation as the pulse width is already defined by thezero current switching criterion. Thus, modulation of the repetitionrate of the pulses is the only way of modulating the currents flowingthrough lines 26 and 27 for providing sine-shaped half waves and tocontrol the power fed from the power generator connected to the inputterminals 2 and 3 into an AC grid connected to the output terminals 4and 5 by the power inverter 1. Thus, the controller 41 varies therepetition rate of the pulses of the signals R and L within each periodof the grid frequency for sine-shaping the currents through the lines 26and 27, and further varies the average value of the repetition rate ofthe pulses in the control signals R and L to optimize the power fed intothe AC grid. This may be done according to a generally known maximumpower point (MPP) tracking method. Additionally, the controller 41optimizes the time shift of the control signals R and L or the overlapof their pulses 63, respectively, for zero current switching thesemiconductor power switches 9 to 12 of the half bridges 7 and 8 at therespective voltage applied between the lines 26 and 27 by the AC gridconnected to the output terminals 4, 5. These various optimizations maybe achieved in that a look-up table is stored in the controller and thatthe controller looks up suitable on-times and repetition rates or evenready-to-use pulse sequences in that table depending on theinstantaneous voltage difference between the output terminals 4 and 5monitored by the controller at Va and Vb. A typical range within whichthe controller varies the repetition of the pulses in the controlsignals R and L extends from 20 to 500 KHz.

In the power inverter 1, an isolation barrier 50 is formed that enclosesall parts from the secondary winding 18 of the high frequencytransformer 17 up to the output terminals 4 and 5. In this isolated partof the power inverter 1 only very little electric energy is stored atany time as the secondary winding 18 is of small inductance and as thecapacitors 20, 21 and 25 are all of small capacitance. Thus, as long asthe half bridges 7 and 8 are not operated, and as long as no grid isconnected to the output terminals 4 and 5, touching any component of thepower inverter enclosed by the isolation barrier is without risk, evenif a power generator is connected to the input terminals 2 and 3 andapplies a voltage between these input terminals. In the direction of thepower flux from the input terminals 2 and 3 to the output terminals 4and 5 the isolation barrier is not only provided by the transformer 17but also by the capacitors 20 and 21 that are providing an additionalgalvanic separation and more or less block currents with the gridfrequency of 50 Hz, for example. In the signal lines 42 and 43 theisolation barrier is provided by the capacitor 46 and by the high ohmicresistors 44 and 45 arranged on both sides of the barrier 50. In theconnection between the input terminal 3 and the line 27 the tworesistors 36 and 37 provide the isolation barrier between them.

In the power inverter 1, a reference voltage Vref from a controllerinternal ADC is used for providing a defined 50% voltage offset by meansof connecting four ohmic resistors 54 of equal resistance as follows:One resistor 54 is connected between the input terminal 3 and the end ofthe signal line 42 connected to the controller 41 at its terminal Va.One resistor 54 is connected between the input terminal 3 and the end ofthe other signal line 43 connected to the controller 41 at its terminalVb. The two other resistors 54 are connected between a reference voltagepoint 55 at which the reference voltage Vref is provided and the ends ofthe signal lines 42 and 43 which are connected to the controller 41.

The semiconductor power switches 9 to 12 of the half bridges 7 and 8 areoperated under a full wave mode zero current switching scheme, and thetopology of the power inverter 1 according to FIGS. 1 and 2 is tuned tothis switching scheme. As a result, the resonant converter 51 accordingto FIGS. 1 and 2 basically acts as a current source, i.e., a powersource that, independently of the voltage applied between these lines 26and 27 by the AC grid connected to the output terminals 4, 5, provides acurrent to the lines 26 and 27, which is shaped at the primary side ofthe high frequency transformer 17. This, however, does not exactly applywhen the instantaneous voltages between the lines 26, 27 are close tozero and when the corresponding currents to be fed into the AC grid arelow. In that case, a larger part of the electric power fed into theprimary winding 16 of the high frequency transformer 17 during the firsthalf wave of the resonance period of the resonant converter 19 is fedback into the buffer capacitor 40 during the second half wave of theresonance period before the on-time of the primary winding 16 ends byshort-circuiting it. As a result, the current effectively fed from thepower generator to the AC grid during one on-time of the primary winding16 is reduced. This is advantageous as the repetition rate of the pulsesin the high frequency control signals may be kept at higher levels withvoltages between the lines 26 and 27 close to zero as it would bepossible without the feedback of a part of the power into the buffercapacitor 40. A high repetition rate of the pulses in the high frequencycontrol signals R, L means a better controllability, and no medium tolow frequency noise, which cannot be filtered away by the filtercapacitor 25, is induced by the switching of the semiconductor powerswitches 9 to 12.

FIGS. 6 to 10 illustrate the time course of the current 64 through theprimary winding 16 and the voltage 65 across the secondary winding 18 ofthe high frequency transformer 17 of the power inverter 1 according toFIGS. 1 and 2 during and following one on-time 66 defined by the timeshift between the control signals R and L according to the full wavemode zero current switching scheme, i.e., the on-time 66 ends when thecurrent 64 becomes zero for the second time after the beginning of theon-time 66 and when the current would essentially remain zero with anongoing on-time 66. The voltage across the buffer capacitor applied tothe primary winding 16 during the on-time 66 is the same (25 Volts) forall FIGS. 6 to 10. With a transformer ratio of 1:16 of the highfrequency transformer, this input voltage corresponds to a maximumoutput voltage of the high frequency transformer of 400 V. The externalvoltage that is presently applied to the output lines 26 and 27, forexample by an AC power grid connected to the output terminals 4 and 5via the output converter 31 according to FIG. 1, however, decreases fromFIG. 6 to FIG. 10. It is about 375 Volts for FIG. 6, about 225 Volts forFIG. 7, about 125 Volts for FIG. 8, about 50 Volts for FIG. 9, and about2 Volts for FIG. 10. Considering the voltage applied to the primarywinding and considering the transformer ratio of the high frequencytransformer, these external voltages correspond to relative voltages of94% (FIG. 6), 56% (FIG. 7), 31% (FIG. 8), 12.5% (FIG. 9), and 0.5% (FIG.10). The absolute instantaneous value of the presently applied externalvoltage can indirectly be seen in FIGS. 6 to 9 from a voltage drop 67occurring at the beginning of the second half wave of the resonanceperiod, when the diodes 24 of the rectifier full bridge 23, which havepreviously been conductive, are commutated due to the changing directionof the current 64. With a high value of the presently applied externalvoltage, the electric power fed into the resonant series circuit 19during the first half wave of its resonance period is transferredthrough the high frequency rectifier 22 into the lines 26 and 27particularly easily. As a result, the current 64 according to FIG. 6drops to low values and essentially remains there already at the end ofthe first half wave of the resonance period, when the diodes 24 of therectifier full bridge 23, which have previously been conductive, arecommutated with the change of direction of the current 64. This allowsfor an early termination of the on-time 66 at a current 64 of zero orclose to zero. The time of the second zero crossing of the current 64determining the end of the on-time 66 according to FIG. 6 is no longerdefined by the resonance period of the resonant series circuit 19 but byother time constants. These other time constants resulting in a shorterresonance period, also define the time course of the current 64 and ofthe voltage 65 after the on-time 66. In FIGS. 7 and 8 the time betweenthe first and the second zero crossing of the current 64 is still muchshorter than 50% of the resonance period of the resonant series circuit19, but there is an increasing current 64 through the short-circuitedprimary winding 16 after the on-time 66 that displays a half wave of aduration of about 50% of the resonance period of the resonant seriescircuit 19. This portion of the current indicates an increasing amountof electric power that is still in the resonant series circuit 19 at theend of the on-time 66 and that is forwarded through the high frequencyrectifier 22 into the lines 26, 27 only after the on-time 66. With afurther decrease of the instantaneous value of the presently appliedexternal voltage the length of the second half wave of the current 64during the on-time 66, i.e., between its first and second zero crossing,gets longer. This behavior becomes predominant at a relative voltage ofabout ⅓ as in FIG. 8; and In FIG. 9 the second half wave has alreadyextended to a length of about 50% of the resonance period of theresonant series circuit 19. As a result, a considerable part of theelectric power fed from the buffer capacitor 40 into the primary winding16 during the first half wave of the current 64 is fed back into thebuffer capacitor 40 during the second half wave, which is accounted forby extending the length of the on-time 66 in FIG. 9. This procedure ofextending the on-time at low instantaneous output voltages and, thus,feeding back energy into the buffer capacitor 40 reduces the electricpower transferred by the high frequency transformer 17 during oneon-time 66 and allows for maintaining a high repetition rate of theon-times 66 even with low instantaneous output voltages, low outputcurrents and corresponding low instantaneous output powers. The electricpower that is still in the resonant series circuit 19 after the on-timeoscillates in the resonant series circuit 19 and is transferred fromthere through the high frequency rectifier 22 into the lines 26 and 27.Thus, the two half waves of the current 64 after the end of the on-time66 seen in FIGS. 9 and 10 indicate that a part of the electric power isstill transferred from the power generator to the AC grid. In FIG. 10this fraction of transferred power is strongly reduced due to the verylow instantaneous output voltage, i.e., only very little electric poweris effectively transferred during the on-time 66 according to FIG. 10while the major part initially transferred during the first half wave isstored back into the buffer capacitor 40 during the second half wave ofthe resonant frequency of the resonant series circuit 19. This allowsfor a still high repetition rate of the on-times 66 at an instantaneousoutput voltage close to zero and with corresponding output current andinstantaneous output power close to zero. While FIGS. 6 to 10 indicatethat the on-times 66 may be terminated by monitoring the current 64 andwaiting for its second zero crossing after the beginning of the on-times66, the on-times 66 are, in one embodiment, selected from a table basedon the instantaneous value of the external voltage, or even better on aratio of this instantaneous value of the external voltage and thepresent value of the voltage across the buffer capacitor applied to theprimary winding during the on-times 66, or on the instantaneous relativevoltage as defined above.

FIG. 11 shows an alternative scheme for the high frequency controlsignals R and L according to FIG. 5. In this scheme, the length of theon-times 66 is not defined by a time shift between the high frequencycontrol signals R, L but by the length of dips 68 in the signals R, Lthat are otherwise on, and that exhibit a fixed time shift between thesignals R and L of half the period of the signals R and L. Thecorresponding inverted control signals R,inv. and L,inv. exhibit pulses69 of the same length as the dips 68. While the primary winding 16 ofthe high frequency transformer 17 according to FIGS. 1 and 2 whenclocked with the scheme according to FIG. 5 is short-circuited in analternating manner by either the semiconductor power switches 9 and 11that are controlled by the inverted control signals R,inv. and L,inv. orby the semiconductor power switches 10 and 12 that are controlled by theoriginal control signals R and L, as shown by cross-hatched intervals inFIG. 5, the high frequency control signals R, L according to FIG. 11result in the primary winding 16 being short-circuited always by thesemiconductor power switches 10 and 12 that are controlled by the highfrequency signals R and L, as shown by cross-hatched intervals in FIG.11. This can be used advantageously to customize the spatial arrangementof the semiconductor power switches 9 to 12 with regard to the primarywinding 16, e.g., in a board layout of the inverter 1, as indicated inFIG. 12. The semiconductor power switches 10 and 12 that are directlycontrolled by the control signals R, L are arranged in FIG. 12 in such away that they connect the center points 13 and 14 and, thus, the ends ofthe primary winding 16 via a shorter current path than the semiconductorpower switches 9 and 11. Thus, electrical losses due to currentscirculating through the short-circuited primary winding 16 after theon-times 66 are kept at a minimum.

Zero current switching of the semiconductor power switches 9 to 12 maynot reduce the switching losses to the same extent as zero voltageswitching. However, this is more than outweighed by the fact that theswitching losses at the limited input voltages of the power inverter 1are also limited, that the controlling losses, i.e., the electric powerneeded for controlling the semiconductor power switches 9 to 12 inoperation, are particularly low with zero current switching, i.e., lowerthan with zero voltage switching, and that only full wave mode zerocurrent switching allows for a reduced power transfer during eachon-time in case of low instantaneous output voltages. This reduced powertransfer in turn allows for keeping a high repetition rate of thepulses, i.e., a high switching frequency, even at these lowinstantaneous output voltages that occur twice during each period of thealternating voltage of the AC grid connected to the output terminals 4,5.

FIG. 13 shows four power inverters 1, as an example for a plurality ofpower inverters 1, which may include much more individual powerinverters 1 that each feed electric energy from a photovoltaic module 70as a power generator 71 via a ring 72 of lines 74 to 76 and a relay 73into the AC grid 77. All output terminals 4 of all power inverters 1 areconnected to the same ring shaped power line 74, all output terminals 5of all power inverters 1 are connected to the ring shaped power line 75,and all ground terminals 6 of all power inverters 1 are connected to thering shaped ground line 76. The ring shape of the lines 74 to 76 hasboth advantages with regard to failure safety and ohmic losses. The ring72 serves as an AC power collecting bus. The AC power collected on thisbus is fed into the grid 77 via the central relay 73 that is arranged ina box 78 separately from all the individual power inverters 1. The powerinverters 1 are arranged close to the photovoltaic modules 70, and mayin fact be mounted to the backsides of the photovoltaic modules 70. Therelay 73 is just a back-up shut-off means for the inverters 1, as eachinverter may completely be shut-off by deactivating its drivers 52, 53for its semiconductor switches 9 to 12 and as each inverter includes agalvanic separation between the AC grid 77 and the photovoltaic module70 due to the high frequency transformer 17 and the isolation barrier50.

Many variations and modifications may be made to the embodiments of thedisclosure without departing substantially from the spirit andprinciples of the disclosure. All such modifications and variations areintended to be included herein within the scope of the presentdisclosure, as defined by the following claims.

What is claimed is:
 1. A power inverter for feeding electric energy froma DC power generator into an AC grid with two power lines, comprising:two input terminals configured to connect to the DC power generator; twooutput terminals configured to connect the two power lines of the ACgrid; a resonant converter comprising: a high frequency transformercomprising a primary winding and a secondary winding, at least one highfrequency switched semiconductor power switch that connects one end ofthe primary winding of the high frequency transformer to one of theinput terminals and provides a current path through the primary windingto the other one of the input terminals, a resonant series circuitcomprising an inductance and a capacitance, and a high frequencyrectifier configured to rectify a current through the secondary windingof the high frequency transformer and having two output lines; and anoutput converter connected between the output lines of the highfrequency rectifier and the two output terminals, wherein the resonantconverter further comprises a controller that is connected to the outputterminals and configured to receive a voltage signal and provide highfrequency control signals for controlling the at least one semiconductorpower switch of the resonant converter to sine-modulate an AC currentfed into the AC grid in phase with the voltage signal.
 2. The powerinverter of claim 1, wherein the controller is configured to vary arepetition rate of pulses in the high frequency control signals forsine-modulating the AC current fed into the AC grid.
 3. The powerinverter of claim 2, wherein the controller is further configured tovary an average repetition rate of the pulses in the high frequencycontrol signals for controlling the electric power fed from the powergenerator into the AC grid.
 4. The power inverter of claim 1, whereinthe resonant converter further comprises two half bridges connectedbetween the input terminals, each half bridge having two high frequencyswitched semiconductor power switches and a center, wherein the primarywinding of the high frequency transformer is connected between thecenters of the two half bridges.
 5. The power inverter of claim 4,wherein the controller is configured to provide the high frequencycontrol signals for controlling the high frequency switchedsemiconductor power switches of the two half bridges in a way such as toeither connect one end of the primary winding of the high frequencytransformer to one of the input terminals and the other end of theprimary winding to the other one of the input terminals, or toshort-circuit the two ends of the primary winding via one high frequencyswitched semiconductor power switch of each of the two half bridges. 6.The power inverter of claim 4, wherein the controller is configured toprovide the two control signals at a time shift with partiallyoverlapping pulses, the partially overlapping pulses in both controlsignals being of equal length that is at least as long as a resonanceperiod of the resonant series circuit of the resonant converter, andwherein the controller is configured to vary the time shift between thetwo high frequency signals for varying the length of the on-times. 7.The power inverter of claim 6, wherein both control signals exhibit aduty cycle of about 50% independent of a repetition rate of the pulses.8. The power inverter of claim 5, wherein the controller is configuredto provide the high frequency control signals for controlling the highfrequency switched semiconductor power switches of the two half bridgesin a way such as to short-circuit the two ends of the primary windingvia always the same two high frequency switched semiconductor powerswitches.
 9. The power inverter of claim 8, wherein a current pathbetween the two ends of the primary winding through the always same twohigh frequency switched semiconductor power switches is configured tohave essentially lower electrical losses than a current path between thetwo ends of the primary winding through the two other high frequencyswitched semiconductor power switches of the two half bridges.
 10. Thepower inverter of claim 4, wherein the controller is configured toprovide one high frequency control signal per half bridge that isdirectly controlling one of the two semiconductor power switches of therespective half bridge and that is inverted for controlling the other ofthe two semiconductor power switches of the respective half bridge. 11.The power inverter of claim 10, wherein the controller is configured toprovide the high frequency control signals for zero current switching ofthe semiconductor power switches of the half bridges.
 12. The powerinverter of claim 11, wherein the controller is configured to providethe high frequency control signals for full wave mode zero currentswitching of the semiconductor power switches of the half bridges,wherein the controller is configured to vary a length of on-times of theprimary winding, during which one of the semiconductor power switches ofone of the half bridges that is connected to one of the input terminalsand one of the semiconductor power switches of the other of the halfbridges that is connected to the other of the input terminals are closedwhile the two other semiconductor power switches of the half bridges areopen, in a range from about 50% of the resonance period at a maximumvoltage signal to about 100% of the resonance period at a zero voltagesignal.
 13. The power inverter of claim 10, wherein the controller isconfigured to modulate the repetition rate of the pulses in the highfrequency control signals and/or vary the length of the on-timesaccording to control data stored in a look-up table depending on thevoltage signal.
 14. The power inverter of claim 4, wherein thecontroller is configured to provide the two high frequency controlsignals to the semiconductor power switches via two drivers each ofwhich is provided for one of the two half bridges and closes one of thesemiconductor power switches of the respective half bridge when theprovided high frequency control signal is high and closes the other ofthe semiconductor power switches of the one half bridge, when theprovided high frequency control signal is low, with the semiconductorpower switches of the two half bridges that are closed by the twodrivers, when both of the two high frequency control signals are high,being connected to the same input terminal.
 15. The power inverter ofclaim 1, wherein the resonant series circuit comprises the highfrequency transformer as the inductance, and two capacitors as thecapacitance, the secondary winding of the high frequency transformer isconnected between the two capacitors, and wherein the high frequencyrectifier is connected to the capacitors.
 16. The power inverter ofclaim 15, wherein the resonant series circuit consists of the twocapacitors of equal capacitance and the high frequency transformer. 17.The power inverter of claim 1, wherein the high frequency rectifiercomprises a rectifier full bridge.
 18. The power inverter of claim 1,wherein the high frequency rectifier comprises a low-pass filtercapacitor.
 19. The power inverter of claim 1, wherein the outputconverter comprises a line-commutated converter commutated via thevoltages applied to the output terminals.
 20. The power inverter ofclaim 19, further comprising a ground terminal configured to connect apower grid ground line, wherein one of the input terminals is connectedto one of the output lines of the high frequency rectifier via aresistor on the one hand, and to the ground terminal via a capacitor onthe other hand.
 21. The power inverter of claim 20, further comprising aresistor connected in parallel to the capacitor connecting the one ofthe input terminals.
 22. The power inverter of claim 20, wherein the oneof the output lines of the high frequency rectifier connected to the oneof the input terminals is the output line of the high frequencyrectifier that exhibits the lower electrical potential of the inputterminals with regard to electric ground.
 23. The power inverter ofclaim 1, further comprising an electrical isolation barrier between theinput terminals and the output terminals, with all high frequencyswitched semiconductor power switches and the primary winding of thehigh frequency transformer being on the same side of the barrier as theinput terminals, and with the resonant series circuit including thesecondary winding of the high frequency transformer, the high frequencyrectifier and the output converter being on the same side of the barrieras the output terminals.
 24. The power inverter of claim 22, wherein alllines crossing the isolation barrier comprise at least one of thefollowing: a high ohmic resistor or a capacitor at the isolationbarrier.
 25. The power inverter of claim 1, further comprising anEMC-filter comprising chokes selected from common mode chokes and combimode chokes connected between the output converter and the two outputterminals, and a varistor connected between taps of the chokes thatcorrespond to each other.
 26. The power inverter of claim 25, whereinthe chokes are combi mode chokes that are arranged on a common ringmagnetic core that has an additional web comprising an air gap in whichan electrically conductive material is provided.
 27. The power inverterof claim 1, wherein the high frequency transformer provides a galvanicseparation between the two input terminals and the two output lines,wherein the output converter comprises a line-commutated convertercommutated via the voltages applied to the output terminals, and furthercomprising a ground terminal configured to connect to a power gridground line, wherein one of the input terminals is connected to one ofthe output lines via a resistor on the one hand, and to the groundterminal via a capacitor on the other hand.
 28. The power inverter ofclaim 27, further comprising a resistor connected in parallel to thecapacitor connecting to one of the input terminals.
 29. The powerinverter of claim 27, wherein the one of the output lines that isconnected to the one of the input terminals is the line that exhibitsthe lower electrical potential with regard to electric ground.
 30. Thepower inverter of claim 27, wherein the one of the output lines that isconnected to the one of the input terminals is the output line thatexhibits the higher electrical potential with regard to electric ground.31. A method of operating the power inverter that comprises: two inputterminals configured to connect to the DC power generator; two outputterminals configured to connect the two power lines of the AC grid; aresonant converter comprising: a high frequency transformer comprising aprimary winding and a secondary winding, at least one high frequencyswitched semiconductor power switch that connects one end of the primarywinding of the high frequency transformer to one of the input terminalsand provides a current path through the primary winding to the other oneof the input terminals, a resonant series circuit comprising aninductance and a capacitance, and a high frequency rectifier configuredto rectify a current through the secondary winding of the high frequencytransformer and having two output lines; and an output converterconnected between the output lines of the high frequency rectifier andthe two output terminals, wherein the resonant converter furthercomprises a controller that is connected to the output terminals andconfigured to receive a voltage signal and provide high frequencycontrol signals for controlling the at least one semiconductor powerswitch of the resonant converter to sine-modulate an AC current fed intothe AC grid in phase with the voltage signal, the method comprising:generating via the controller two high frequency control signals at atime shift between the two high frequency control signals, and applyingthe two high frequency control signals to those two semiconductor powerswitches of the two half bridges that are connected to one of the twoinput terminals to provide for on-times during which only one of thesetwo semiconductor power switches is closed; and generating two inversehigh frequency control signals each being the inverse of one of the twohigh frequency control signals, and applying the two inverse highfrequency control signals to those two semiconductor power switches ofthe two half bridges that are connected to the other of the two inputterminals such that each high frequency control signal and the inverseof that control signal are applied to the respective two semiconductorpower switches of the same half bridge; and synchronously modulating allthe high frequency control signals to sine-modulate the AC current fedinto the AC grid in phase with a voltage of the AC grid.
 32. The methodof claim 31, further comprising varying a repetition rate of pulses inthe high frequency control signals for sine-modulating the AC currentfed into the AC grid.
 33. The method of claim 32, further comprisingvarying an average repetition rate of the pulses in the high frequencycontrol signals for controlling the electric power fed from aphotovoltaic generator into the AC grid.
 34. The method of claim 31,further comprising varying a length of the on-times in a range fromabout 50% to about 100% of a resonance period of the resonant seriescircuit of the resonant converter for full wave mode zero currentswitching of the semiconductor power switches of the half bridges. 35.The method of claim 34, wherein the length of the on-times is variedbased on a present ratio of an external voltage applied to the outputlines of the high frequency rectifier and a present value of a voltagethat is applied to the primary winding of the high frequency transformerduring the on-times.
 36. The method of claim 31, wherein both controlsignals are generated by the controller by partially overlapping pulseswith a duty cycle of about 50% independent of the repetition rate of thepulses, and wherein the time shift between the control signals is variedfor varying the length of the on-times.